Semiconductor integrated circuit device having power reduction mechanism

ABSTRACT

A semiconductor integrated circuit device is composed of logic gates each provided with at least two MOS transistors. The logic gates are connected to a first potential point and a second potential point. The semiconductor integrated circuit device includes a current control device connected between the logic gate and the first potential point and/or between the logic gate and the second potential point for controlling a value of a current flowing in the logic gate depending on an operating state of the logic gate. The circuit can be used in devices that cycle in operation between high and low power consumption modes, such as microprocessors that have both an operation mode and a low power back-up or sleep mode used for power reduction.

This is a continuation application of U.S. Ser. No. 08/653,248 filed May24, 1996, now U.S. Pat. No. 5,828,235 which is a continuationapplication of Ser. No. 08/193,765 filed Feb. 8, 1994, now U.S. Pat. No.5,583,457 which is a Continuation-In-Part of U.S. patent applicationSer. No. 08/045,792 filed Apr. 14, 1993.

BACKGROUND OF THE INVENTION

The present invention relates to a semiconductor integrated circuitcomposed of scaled MOS transistors, and more particularly to a circuitsuitable for high-speed and low power operation and an electronic deviceusing the same.

As the size of MOS transistors is scaled down, the breakdown voltagethereof is lowered as stated, for example, in the 1989 InternationalSymposium on VLSI Technology, Systems and Applications, Proceedings ofTechnical Papers, pp. 188-192 (May 1989). Accordingly, the operatingvoltage thereof has to be lowered. In particular, the operating voltageis lowered even more for the purpose of achieving low power consumptionfor semiconductor devices used in a battery-operated portable equipmentand the like.

SUMMARY OF THE INVENTION

It is an object of the present invention to provide a semiconductorintegrated circuit capable of operating at high speed and with low powerconsumption even when the size of MOS transistors is scaled down.

It is another object of the present invention to provide an electronicdevice capable of operating at high speed and with low power consumptionand suitable for being battery powered drive even when the size of MOStransistors is scaled down.

To decrease power consumption for an integrated circuit having MOStransistors, it is necessary to lower the threshold voltage V_(T) of thetransistor in accord with the lowering of the operating voltage in orderto maintain a high-speed operation. This is due to the fact that theoperating speed is governed by the effective gate voltage of the MOStransistor, i.e., a value obtained by subtracting V_(T) from theoperating voltage, and the larger this value becomes, the higher thespeed becomes. For example, a typical value of a threshold voltage of atransistor having a channel length of 0.25 μm and operating at 1.5 V isanticipated to be 0.35 V according to the above-mentioned document.According to a well-known scaling law, the typical value of thethreshold voltage becomes approximately 0.24 V when it is assumed thatthe operating voltage is 1 V. If V_(T) is brought down to approximately0.4 V or lower, however, it becomes no longer possible to turn thetransistor completely off and a D.C. current starts to flow through itdue to the sub-threshold characteristics (tailing characteristics) ofthe MOS transistor as described hereafter. Thus, this current becomes aserious issue in the practical operation of a device having MOStransistors at 1.5 V or lower.

A conventional CMOS inverter shown in FIG. 49 will be described.Ideally, an N-channel MOS transistor M_(N) is turned off when an inputsignal IN is at a low level (=V_(SS)), and a P-channel MOS transistorM_(P) is turned off when IN is at a high level (=V_(CC)), thus nocurrent flows in either case. When V_(T) of the MOS transistor becomeslow, however, the subthreshold current can no longer be disregarded.

As shown in FIG. 50, a drain current I_(DS) in a subthreshold region isin proportion to an exponential function of a gate-source voltage V_(GS)and is expressed by the following expression. ##EQU1## Where, Windicates a channel width of the MOS transistor, I_(O) and W_(O)indicate a current value and a channel width when V_(T) is defined, andS indicates a subthreshold swing (the gate-voltage swing needed toreduce the current by one decade). Thus, a subthreshold current:##EQU2## flows even when V_(GS) =0. Since V_(GS) =0 in the transistor inan off-state of the CMOS inverter shown in FIG. 49, the current I_(L)mentioned above will flow from the high power supply voltage V_(CC)toward the low power supply voltage V_(SS) which is at ground potential,even at the time of non-operation.

This subthreshold current increases exponentially from I_(L) to I_(L) 'when the threshold voltage is lowered from V_(T) to V_(T) ' as shown inFIG. 50.

As is apparent from the above expression (2), it is sufficient either toincrease V_(T) or to decrease S in order to reduce the subthresholdcurrent. However, the former method brings about a lowering of the speeddue to a lowering of the effective gate voltage. In particular, when theoperating voltage is lowered with the scale-down of the breakdownvoltage, the decrease in speed becomes notable and the advantages ofscaled down fabrication can no longer be put to practical use, which isnot preferable. Further, the latter method is difficult to apply forroom temperature operation because of the following reasons.

The subthreshold swing S is expressed by the capacitance C_(OX) of agate insulator and the capacitance C_(D) of a depletion layer under thegate as follows. ##EQU3## Where, k indicates the Boltzmann constant, Tindicates absolute temperature, and q indicate the elementary charge. Asis apparent from the above expression, S≧kT 1n 10/q irrespective ofC_(OX) and C_(D), thus it is difficult to bring it to 60 mV or lower atroom temperature.

The substantial D.C. current of a semiconductor integrated circuitcomposed of a plurality of MOS transistors increases remarkably due tothe phenomenon described above. Namely, since V_(T) has to be made loweras the operating voltage is lowered at a constant operating speed, thesituation becomes more serious when the operation is performed at alower voltage. At the time of operation at a high temperature inparticular, V_(T) becomes lower and S becomes larger. Therefore, thisproblem becomes even more serious. In the times of downsizing ofcomputers or the like for the future when low power consumption isimportant, the increase of the subthreshold current is a substantialissue. In particular, in an electronic device which is desired to beoperated by one power cell of a level of 0.9 to 1.6 V, it is also veryimportant to cope with the increase of the current.

In order to solve the above-described problems, according to the presentinvention, control circuit means for controlling the supply of a largecurrent and a small current is inserted between the source of a MOStransistor and the power supply so as to apply a current to the MOStransistor circuit by switching these currents in accordance with theiruse. For example, a large current is supplied when high-speed operationis required, and a small current is supplied when low power consumptionis required.

Since high-speed operation is required at time of normal operation, alarge current is supplied to the MOS transistor circuit from the currentsupply means so as to make high-speed operation possible. At this time,a D.C. current flows in the MOS transistor circuit as describedpreviously, which, however, is sufficiently small normally as comparedwith the operating current, i.e., charging and discharging current of aload, thus causing no problem.

On the other hand, since low power consumption is required at the timeof standby, the supplied current is changed over to a small current soas to restrain the subthreshold current. At this time, a logic voltageswing of a MOS transistor circuit generally may become smaller than thatat the time of supplying a large current because the current is limited,but there is no problem in so far as ensuring the logic level.

As described above, it is possible to realize a high-speed and low powerconsuming MOS transistor circuit and an electronic device composed ofthe same according to the present invention.

Besides, the present invention has been described with a MOSsemiconductor integrated circuit device as an example, but the presentinvention is applicable to a metal insulator semiconductor (MIS)integrated circuit device in general.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1A is a diagram showing an inverter in an embodiment 1 of thepresent invention;

FIG. 1B is a diagram showing the voltage levels of signals of theinverter in the embodiment 1;

FIG. 2 is a diagram showing a principle of subthreshold currentreduction according to the present invention.

FIG. 3 is a diagram showing the subthreshold current reduction effectaccording to the present invention;

FIG. 4 is a circuit diagram showing an inverter in an embodiment 2 ofthe present invention;

FIGS. 5A to 5C are diagrams showing the timing of signals of the presentinvention;

FIG. 6 is a diagram showing a device structure of the present invention;

FIG. 7 is a circuit diagram of an inverter in an embodiment 3 of thepresent invention;

FIG. 8 is a circuit diagram of an inverter in an embodiment 4 of thepresent invention;

FIG. 9 is a diagram showing a device structure of the present invention;

FIG. 10A is a diagram showing an inverter chain in an embodiment 5 ofthe present invention;

FIG. 10B is a diagram showing voltage levels of signals of the inverterchain in the embodiment 5;

FIG. 11A is a diagram showing an inverter chain in an embodiment 6 ofthe present invention;

FIG. 11B is a diagram showing voltage levels of signals of the inverterchain in the embodiment 6;

FIG. 12A is a diagram showing an inverter chain in an embodiment 7 ofthe present invention;

FIG. 12B is a diagram showing voltage levels of signals of the inverterchain in the embodiment 7;

FIG. 13 is a diagram showing an example of a grouping of a combinationallogic circuit applied with the present invention;

FIG. 14 is a diagram showing a combinational logic circuit in anembodiment 8 of the present invention;

FIG. 15 is a diagram showing a combinational logic circuit in anembodiment 9 of the present invention;

FIGS. 16A and 16B are diagrams showing a latch in an embodiment of thepresent invention;

FIG. 17 is a circuit diagram showing a latch in an embodiment 11 of thepresent invention;

FIG. 18 is a circuit diagram of an inverter chain in an embodiment 12 ofthe present invention;

FIG. 19 is a circuit diagram of an inverter chain in an embodiment 13 ofthe present invention;

FIG. 20 is a circuit diagram of a NAND gate in an embodiment 14 of thepresent invention;

FIG. 21 is a circuit diagram of a NOR gate in an embodiment 15 of thepresent invention;

FIG. 22 is a circuit diagram of a clocked inverter in an embodiment 16of the present invention;

FIG. 23 is a circuit diagram of a combinational logic circuit in anembodiment 17 of the present invention;

FIG. 24 is a circuit diagram of a latch in an embodiment 18 of thepresent invention;

FIG. 25 is a circuit diagram of an output buffer in an embodiment 19 ofthe present invention;

FIG. 26 is a circuit diagram of an input buffer in an embodiment 20 ofthe present invention;

FIG. 27 is a circuit diagram of an NMOS dynamic circuit in an embodiment21 of the present invention;

FIG. 28 is a diagram showing an embodiment 22 of the present invention,conceptually;

FIG. 29 is a circuit diagram of a CMOS inverter in an embodiment 23;

FIG. 30 is an operation timing diagram of a CMOS inverter in anembodiment 23;

FIG. 31 is a diagram showing an inverter chain in an embodiment 24;

FIG. 32 is a diagram showing an inverter chain in an embodiment 25;

FIG. 33 is a diagram showing a CMOS inverter in an embodiment 26;

FIG. 34 is a circuit diagram of a level hold circuit in an embodiment27;

FIG. 35 is a circuit diagram showing a latch capable of providing afixed output;

FIG. 36 is a timing chart for explaining the timing of the operation ofthe circuit of FIG. 35;

FIG. 37 is a circuit diagram of a latch capable of providing a fixedoutput;

FIG. 38 is a timing chart for explaining the timing of the operation ofthe circuit of FIG. 37;

FIG. 39 is a diagram showing a dual-phase clock logic circuit;

FIG. 40 is a circuit diagram showing an inverter operating with adual-phase clock;

FIG. 41 is a timing chart useful for explaining the operation of thecircuit shown in FIGS. 39 and 40.

FIG. 42(a) shows an embodiment of the invention applied to a gate array;

FIG. 42(b) shows a logic diagram;

FIG. 43(a) shows another embodiment of the invention applied to a gatearray;

FIG. 43(b) shows a logic diagram;

FIG. 44 is a block diagram showing a single-chip microprocessorconstructed according to an embodiment of the invention;

FIG. 45 is a diagram showing an internal structure of the co-processorof the microprocessor of FIG. 44;

FIG. 46 is a diagram showing the internal structure of the local memoryof the microprocessor shown in FIG. 44;

FIG. 47 is a diagram showing the internal structure of the bus controlunit for the microprocessor of FIG. 44;

FIG. 48 is a timing diagram useful for explaining the operation of themicroprocessor of FIG. 44, according to the present invention;

FIG. 49 is a circuit diagram of a conventional CMOS inverter; and

FIG. 50 is a diagram showing subthreshold characteristics of a MOStransistor.

DESCRIPTION OF THE PREFERRED EMBODIMENTS

Specific embodiments of the present invention will be described in moredetail hereinafter with reference to the drawings.

Embodiment 1

First, FIGS. 1A and 1B show an embodiment suitable for explaining aprinciple of the present invention.

FIG. 1A is a circuit diagram of an inverter according to an embodimentof the present invention. In FIG. 1A, L represents a CMOS inverter,which is composed of a P-channel MOS transistor M_(P) and an N-channelMOS transistor M_(N). The present invention is applicable not only to aninverter, but also to a logic gates such as NAND and NOR circuits or toa logic gate group as described later. However, a case of an inverterwill be described herein for the sake of simplicity. S_(C) and S_(S)represent switches and R_(C) and R_(S) represent resistances. In thepresent embodiment, switches S_(C) and S_(S) and resistances R_(C) andR_(S) are inserted in parallel between power supply terminals V_(CL) andV_(SL) of an inverter L and power supplies V_(CC) and V_(SS),respectively. With this arrangement, subthreshold current reduction isrealized as described hereinafter.

In a period where high-speed operation is required, the switches S_(C)and S_(S) are switched on, and V_(CC) and V_(SS) are applied directly tothe inverter L (hereinafter referred to as a high-speed operation mode).High-speed operation can be performed if threshold voltages V_(T) ofM_(P) and M_(N) are set low. At this time, a subthreshold current flowsin the inverter L as described previously, which, however, causes noproblem since the current is normally sufficiently small as comparedwith an operating current, i.e., charging and discharging current of aload.

On the other hand, in a period where low power consumption is required,the switches S_(C) and S_(S) are switched off and power is supplied tothe inverter through the resistances R_(C) and R_(S) (hereinafterreferred to as a low power consumption mode). As a result of the voltagedrop due to the fact that the subthreshold current flows throughresistances, V_(CL) drops lower than V_(CC) and V_(SL) rises higher thanV_(SS). As shown in FIG. 2, the subthreshold current decreases due tothe voltage drop by means of the following two types of mechanisms.Incidentally, the following description is given for M_(N) when an inputsignal IN is at a low level (V_(SS)), but the same description appliesto M_(P) when IN is at a high level (V_(CC)).

(i) Since the source potential V_(SL) rises, backgate bias V_(BS)=V_(SS) -V_(SL) =-V_(M) is applied, and the threshold voltage rises fromV_(T0) to V_(T1). The rise portion of the threshold voltage is expressedas: ##EQU4## With this, the subthreshold current decreases from I_(L0)to I_(L1). The rate of decrease is: ##EQU5## Where, K is a body effectfactor. For example, when V_(M) =0.3 V, K=0.4 √V, S=100 mV/decade and2Ψ=0.64 V, the subthreshold current is reduced to 21%.

(ii) Since a source potential V_(SL) rises, the gate-source voltageV_(GS) =V_(SS) -V_(SL) =-V_(M) becomes negative. With this, thesubthreshold current decreases further from I_(L1) to I_(L2). The rateof decrease is: ##EQU6## For example, when V_(M) =0.3 V and S=100mV/decade, the subthreshold current is reduced to 0.1%.

When the effects of (i) and (ii) are put together, the followingexpression is obtained. ##EQU7## For example, when V_(M) =0.3 V, thesubthreshold current is reduced to 0.02%. Where, V_(M) represents thesolution of the following equation. ##EQU8##

Incidentally, the back gates of the MOS transistors M_(P) and M_(N) ofthe inverter L may be connected to respective sources V_(CL) and V_(SL),but it is more desirable to connect the back gates to V_(CC) and V_(SS)as shown in FIG. 1A in order to obtain the effect (i) identified above.

FIG. 3 shows a subthreshold current reduction effect. Here, a futurevery large scale LSI operating at an ultra low voltage is assumed, andcomputation is made with respect to a case that a threshold voltageV_(T0) =0.05 to 0.15 V when the back gate bias is 0 and the sum total Wof the channel widths of transistors in an off state in the whole LSI isW=100 m. The more the resistance is increased, the larger V_(M) becomes,thus increasing the effect. As an extreme case, it is also possible tomake the resistance infinite, i.e., to remove the resistance.

However, as shown in FIG. 1B, the logic voltage swing of the outputsignal OUT becomes smaller than the logic voltage swing of the inputsignal IN. Accordingly, consideration of voltage levels of the signalsin the case of a multistage connection must be taken into account andsuch a case will be described later.

Further, the present invention has a function of compensatingfluctuation of the threshold voltage automatically. Namely, when thethreshold voltage is low and the subthreshold current is large, thevoltage drop V_(M) becomes larger, and, when the threshold voltage ishigh and the subthreshold current is small, V_(M) becomes smaller. Ineither case, the fluctuation of the current is restrained. As isapparent from FIG. 3, the fluctuation of the subthreshold current issmaller as the resistance value becomes larger. For example, when theresistance value is set to 3 KΩ or more, the fluctuation of thesubthreshold current I_(L) is restrained within ±20% even if thethreshold voltage fluctuates by ±0.05 V.

Embodiment 2

Next, a specific method for realizing a switch and a resistanceexplained in the embodiment 1 will be described. FIG. 4 shows an examplein which both the switch and the resistance are realized by MOStransistors.

Switching MOS transistors M_(C1) and M_(S1) are those that have largeconductance, and correspond to the switches S_(C) and S_(S) shown inFIG. 1A, respectively. In a high-speed operation mode, M_(C1) and M_(S1)are turned on by bringing a signal φ_(C) to a low level and φ_(S) to ahigh level. The voltage levels of φ_(C) and φ_(S) may be V_(SS) andV_(CC), respectively, but it may also be arranged so that φ_(C) is setlower than V_(SS) and φ_(S) is set higher than V_(CC) in order to makethe conductances of M_(C1) and M_(S1) larger. It is sufficient to applythe voltages for the above from the outside of a chip or to generate thesame by an on-chip booster circuit well known in an EEPROM and a DRAM.

In a low power consumption mode, M_(C1) and M_(S1) are turned off bybringing φ_(C) to a high level and φ_(S) to a low level conversely tothe above. At this time, the current is cut off by one of two methods,for example. In a first method, φ_(C) is made lower than V_(SS) by meansof the external voltage or the on-chip booster circuit. In a secondmethod, transistors having threshold voltages higher (more enhanced)than those used in the inverter L are used as M_(C1) and M_(S1). Thefirst method has such a merit that the process for producing transistorshaving different threshold voltages is unnecessary. On the other hand,the second method is advantageous since the implementation area does notrequire terminals for receiving external voltages or an on-chip boostercircuit.

MOS transistors M_(C2) and M_(S2) are those that have small conductance,and correspond to the resistances R_(C) and R_(S) shown in FIG. 1A,respectively. These transistors are connected to V_(SS) and V_(CC) atthe gate thereof, respectively, and are always in an on-state. Since itis not required to turn off these transistors, there is no problem evenif the threshold voltages thereof are low.

Incidentally, it is also possible to use an N-channel MOS transistor asM_(C2) and a P-channel MOS transistor as M_(S2). For example, taking anN-channel MOS transistor of M_(C2) as an example, a resistance can berealized effectively by what is called a diode connection in which aterminal connected with the gate and the drain thereof is connected tothe terminal V_(CC) , and the source thereof is connected to theterminal V_(CL). By controlling the channel width and the thresholdvoltage of the N-channel MOS transistor, the voltage V_(CL) can be setto a voltage dropped from V_(CC) by the threshold voltage of theN-channel MOS transistor at the time of standby for instance, thusmaking it possible to reduce the subthreshold current with a largemargin.

Next, an example of an application of the timing to be used with thepresent invention will be described. FIGS. 5A to 5C show examples of thetiming of signals φ_(C) and φ_(S).

FIG. 5A and FIG. 5B show applications where the present invention isapplied to a memory LSI. The memory LSI is brought into an operatingstate when a chip enable signal CE (a complementary signal), which is aclock signal from the outside, is at a low level, and a standby statewhen the signal CE is at a high level. In the case of FIG. 5A, aninternal signal φ_(C) is brought to a low level synchronously with thefall of CE and brought to a high level slightly behind the rise of CE.The internal signal φ_(S) is opposite thereto. Thus, the period shown ata in the figure becomes the high-speed operation mode, and the period bbecomes the low power consumption mode. Generally, in a memory unitusing a plurality of memory LSIs, LSIs in the operating state are smallin number, and a large majority of LSIs are in the standby state.Accordingly, when those LSIs that are in the standby state are broughtinto a low power consumption state using the present invention, itcontributes greatly to obtaining a low power consumption of a memoryunit. Besides, the reason for providing delay from the rise of CE untilentering into the low power consumption mode is that an internal circuitof the LSI is reset in the interim.

FIG. 5B shows an example in which further low power consumption isachieved. Here, only immediately after CE has changed is the LSIoperation period brought into the high-speed operation mode. Namely,read-write of data are performed immediately after CE is brought to alow level, and the internal circuit is reset immediately after CE isbrought to a high level. Therefore, these LSI operation periods arebrought into the high-speed operation mode according to the presentinvention, and other LSI operation periods are brought into the lowpower consumption mode according to the present invention.Alternatively, the high-speed operation mode may be entered into when anaddress signal changes.

FIG. 5C shows an example in which the present invention is applied to amicroprocessor. A clock signal CLK is applied in a normal operationstate. At this time, the signal φ_(C) is at a low level and φ_(S) is ata high level, presenting the high-speed operation mode. When themicroprocessor is brought into a standby state or a data holding state,the clock signal CLK is suspended, and a signal BU is brought to a highlevel. φ_(C) shows the high level and φ_(S) shows the low levelsynchronously with the above, presenting the low power consumption mode.With this, the power consumption of the microprocessor is reduced, thusmaking it possible to operate the microprocessor in a backup mode for along period of time with a power supply of small capacity, such as abattery.

FIG. 6 shows an example of a device having a semiconductor structure forrealizing the circuit shown in FIG. 4. Polysilicon 130, 131, 132 and 133in the figure correspond to gates of M_(C2), M_(P), M_(N) and M_(S2)shown in FIG. 4, respectively (M_(C1) and M_(S1) are not shown here).

It is to be noted that M_(C2) and M_(P) hold a same n-well 101 which isconnected to V_(CC) through an n+ diffusion layer 120 in common. M_(N)and M_(S2) also hold a p-substrate connected to V_(SS) in common in asimilar manner as above. As it is understood from the foregoing, it notonly produces the effect (i) described heretofore, but also produces amore advantageous layout area to connect the back gates of the MOStransistors to V_(CC) and V_(SS) as compared with connection of the sameto the sources.

Although an n-well is formed in a p-substrate in the example shown here,a p-well may also be formed in an n-substrate conversely to the above.Otherwise, a triple well structure such as described in the ISSCC Digestof Technical Papers, pp. 248-249, Feb. 1989 may also be adopted.

Embodiment 3

FIG. 7 shows another method for realizing a switch and a resistance. Thefeature of the present embodiment is to use a current mirror circuit.Namely, MOS transistors M_(C2) and M_(C3) having the same thresholdvoltage form what is called a current mirror circuit, and when a currentin proportion to a current source I₀ flows in M_(C2) the impedancethereof is large. The same is also applied to M_(S2) and M_(S3). Thus,M_(C2) and M_(S2) may be regarded as having high resistance. Besides, acircuit CS composed of the current source I₀ and M_(C3) and M_(S3) maybe held in common by a plurality of logic gates.

The current mirror circuit is not limited to the circuit illustratedthere, but also another circuit may be adopted. For example, a bipolartransistor may be used in place of the MOS transistor.

As described, various modifications are possible for the method forrealizing a switch and a resistance. In a word, any means for applying alarge current in a period where high-speed operation is required and forapplying a small current in a period where low power consumption isrequired will suffice. The drawings will be illustrated with switchesand resistances hereinafter as shown in FIG. 1 for the sake ofsimplicity.

Embodiment 4

The backgates of the MOS transistors of the inverter may be connected toother power supplies not limiting to V_(CC) and V_(SS), and the voltagesthereof may also be made variable. An example is shown in FIG. 8. Theback gates of M_(P) and M_(N) are connected here to power suppliesV_(WW) and V_(BB), respectively, and backgate voltage values thereof arechanged depending on time of operation and time of standby. As toV_(BB), in the period where high-speed operation is required, V_(BB) ismade shallow (or slightly positive in an extreme case), and V_(T) ofM_(N) is lowered so as to make high-speed operation possible. In theperiod where low power consumption is required, V_(BB) is made deep andV_(T) of M_(N) is raised, thereby to restrain the subthreshold current.With this, the effect (i) described previously is increased further.V_(BB) has been described above, but the same is applied to V_(WW)except that the polarity of the voltage is reversed. Incidentally, aback gate voltage generating circuit of this sort is described in ISSCCDigest of Technical Papers, pp. 254-255, Feb. 1985.

FIG. 9 shows an example of a device structure for realizing the circuitshown in FIG. 8. The triple well structure described previously is usedhere, in which an n-well 105 (a backgate of the P-channel MOStransistors) is connected to V_(WW) through an n+ diffusion layer 120,and a p-well well 103 (a backgate of the N-channel MOS transistors) isconnected to V_(BB) through a p+ diffusion layer 127.

This triple well structure has such an advantage that the backgatevoltage can be set for every circuit because both the P-channel and theN-channel transistors can be incorporated in respective wells for everycircuit. For example, when a circuit in an operating state and a circuitin a standby state are included in one LSI, it is possible to make thebackgate voltage of the former shallow and the backgate voltage of thelatter deep.

Embodiment 5

Next, a case of an inverter chain in which inverters are connectedsuccessively will be described. The principle will be described withrespect to a case of two stages first for the sake of simplicity.

FIG. 10A shows a circuit diagram for CMOS inverters L₁ and L₂. SwitchesS_(Ci) and S_(Si) and resistances R_(Ci) and R_(Si) (i=1,2) are providedfor each inverter at every stage.

In the high speed operation mode, all four switches are switched on, andV_(CC) and V_(SS) are applied directly to the inverters L₁ and L₂.High-speed operation is made possible by setting the threshold voltagesof the MOS transistors of the inverters low. On the other hand, in thelow power consumption mode, all of the switches are switched off, andthe power is supplied to the inverters through resistances. VoltagesV_(CL1) and V_(CL2) fall lower than V_(CC) , and voltages V_(SL1) andV_(SL2) rise higher than V_(SS) by voltage drops due to the fact thatthe subthreshold current flows through the resistances.

As to the inverter L₁ at the first stage, the subthreshold currentdecreases by the mechanisms described previously (effects (i) and (ii))in a similar manner as to the case of FIG. 1. However, the logic voltageswing of the output N₁ of L₁ is smaller than the logic voltage swing ofthe input signal IN. Namely, the voltage level of N₁ shows V_(CL1) whenIN is at a low level (=V_(SS)), and the voltage level of N₁ showsV_(SL1) when IN is at a high level (=V_(CC)). Since N₁ is the input tothe inverter L₂ at the second stage, it is desirable to set theresistance values so that V_(CC) >V_(CL1) >V_(CL2) and V_(SS) <V_(SL1)<V_(SL2) are effected for the subthreshold current reduction of L₂. Withthis, the subthreshold current decreases by the mechanisms describedpreviously (effects (i) and (ii)) with respect to L₂, too. When V_(CL1)=V_(CL2) and V_(SL1) =V_(SL2), the effect (i) is obtainable, but theeffect (ii) is not obtainable.

Embodiment 6

The same embodiment is also applied to a multistage connection shown inFIG. 11A, and it is recommended to effect V_(CC) >V_(CL1) >V_(CL2) > . .. >V_(CLK) and V_(SS) <V_(SL1) <V_(SL2) < . . . <V_(SLK). Since thelogic voltage swing becomes smaller step by step as shown in FIG. 11B,however, the voltage swing is recovered by inserting a level conversioncircuit appropriately. In the present example, a level conversioncircuit LC is added after the inverter at a Kth stage so that the logicvoltage swing of an output signal OUT becomes the same as that of aninput signal IN. A level conversion circuit of this sort is described inSymposium on VLSI Circuits, Digest of Technical Papers, pp. 82-83, June1992 for instance.

The level conversion circuit LC is not required at the time ofhigh-speed operation. The reason is that, since all the switches are inan on-state, V_(CL1) =V_(CL2) = . . . =V_(CLK) =V_(CC) and V_(SL1)=V_(SL2) = . . . =V_(SLK) =V_(SS) and the logic voltage swing is notreduced. Thus, it is possible to avoid the delay by switching the switchS_(LC) on so as to bypass the level conversion circuit at the time ofhigh-speed operation.

Embodiment 7

FIG. 12A shows another example of an inverter chain of multistageconnection. In the present example, all of switches S_(C) and S_(S) andresistances R_(C) and R_(S) are held in common by means of the invertersL₁ to L_(K), and the voltages V_(CL) and V_(SL) are common to L₁ toL_(K). Therefore, the subthreshold current reduction effect (i)described previously is obtainable, but the effect (ii) is notobtainable as described with reference to FIG. 10. Thus, thesubthreshold current reduction effect becomes smaller than that in theprevious embodiment.

On the other hand, however, there is such an advantage that the layoutarea of switches and resistances can be saved. Further, there is such afeature that the voltage levels of all the signals includinginput-output signals are the same and there is no reduction in the logicvoltage swing as in the previous embodiment as shown in FIG. 12B. As aresult, there is such a merit that the level conversion circuit is notrequired, and a logic circuit such as a NAND circuit, a NOR circuit orthe like is fabricated easily.

Embodiment 8

Next, a case where the present invention is applied to a generalcombinational logic circuit will be described.

For example, a combinational logic circuit shown in FIG. 13 isconsidered. In order to apply the present invention thereto, logic gatesare grouped first as shown in FIG. 13. In the present example, 15 piecesof logic gates L₁ to L₁₅ are divided into three groups G₁, G₂ and G₃. Ingrouping, it is arranged so that the output signals of logic gatesincluded in the (i)th group are inputted only to logic gates of the(i+1)th group and thereafter.

Next, switches and resistances are inserted between each logic gategroup and the power supplies as shown in FIG. 14. Since the logicvoltage swing of the output signal of the logic gate becomes smallerstep by step similarly to the case shown in FIG. 11B, level conversioncircuit groups GC₁ and GC₂ are inserted as shown in FIG. 14 so as torecover the voltage swing. Besides, although it is not illustrated, thelevel conversion circuit groups GC₁ and GC₂ may be bypassed at time ofhigh-speed operation similarly to the case of FIG. 11A.

One of the features of the present embodiment is that logic gatesincluded in the same group hold the switch and the resistance in common.Speaking of the example shown in FIG. 13, three inverters included inthe group G₁ hold the switches S_(C1) and S_(S1) and the resistancesR_(C1) and R_(S1) in common.

Another feature of the present embodiment is that the switch and theresistance are held in common by groups before and after the levelconversion circuit. Namely, groups G₁ and G_(K+1) hold the switchesS_(C1) and S_(S1) and the resistances R_(C1) and R_(S1) in common,groups G₂ and G_(K+2) hold the switches S_(C2) and S_(S2) and theresistances R_(C2) and R_(S2) in common, and groups G_(K) and G_(2K)hold the switches S_(CK) and S_(SK) and the resistances R_(CK) andR_(SK) in common, respectively.

It is possible to reduce the number of switches and resistances for thewhole LSI so as to reduce the layout area by holding the switches andthe resistances in common by a plurality of logic gates as describedabove.

Embodiment 9

FIG. 15 shows another embodiment of the present invention. What differsfrom embodiments described up to this point in the embodiment shown inFIG. 15 is that voltage limiters (voltage down converters and voltage upconverters) VC₁, VC₂, . . . , VC_(K) and VS₁, VS₂, . . . , VS_(K) areused.

When low power consumption is required, switches T_(C1) to T_(CK) andT_(S1) to T_(SK) are changed over to the illustrated sides, and thepower is supplied to the logic gate groups by means of voltage limiters.The voltage limiters VC₁, VC₂, . . . , VC_(K) operate as voltage downconverters on the side of the power supply voltage V_(CC), and generatealmost stabilized internal voltages V_(CL1), V_(CL2), . . . , V_(CLK)lower than V_(CC), respectively. On the other hand, VS₁, VS₂, . . . ,VS_(K) operate as voltage up converters on the side of ground V_(SS),and generate almost stabilized internal voltage V_(SL1), V_(SL2), . . ., V_(SLK) higher than V_(SS), respectively. It is recommended to effectV_(CC) >V_(CL1) >V_(CL) 2 > . . . >V_(CLK) and V_(SS) <V_(SL1) <V_(SL2)< . . . <V_(SLK) for the generated voltages similarly to the embodimentdescribed previously. Incidentally, a voltage limiter of this sort hasbeen disclosed in JP-A-2-246516.

In contrast with the above, when high-speed operation is required, theswitches are changed over to the side opposite to that which isillustrated and V_(CC) and V_(SS) are applied directly to the logic gategroups, thus making high-speed operation possible. Besides, since thevoltage limiters become unnecessary at this time, the operation may besuspended.

Embodiments 10, 11

The circuits without feedback such as an inverter chain and acombinational logic circuit have been used in the embodiments up to thispoint, but the present invention is also applicable to a circuit withfeedback. A case of a latch circuit obtained by combining two NAND gatesshown in FIG. 16A will be described as an example.

FIG. 16B shows a circuit diagram. Switches S_(C1), S_(S1), S_(C2) andS_(S2) and resistances R_(C1), R_(S1), R_(C2) and R_(S2) are insertedamong two NAND gates L₁ and L₂, the power supply V_(CC) and the groundV_(SS), respectively. V_(CL1) and V_(CL2) fall lower than V_(CC),V_(SL1) and V_(SL2) rise higher than V_(SS), and the subthresholdcurrent is reduced by the effect (i) described previously.

FIG. 17 shows an example in which the threshold voltage V_(T) of fourMOS transistors M_(P12), M_(P22), M_(N12) and M_(N22) used for latchinginformation is made higher (more enhanced) than the threshold voltage ofother MOS transistors M_(P11), M_(P21), M_(N11) and M_(N21) in order tofurther reduce the subthreshold current. Since the threshold voltageV_(T) of other MOS transistors M_(P11) M_(P21), M_(N11) and M_(N21) towhich the input signal is applied is left as it is (low), high-speedoperation is possible. In this case, switches and resistances on theV_(SS) side are not required because it is possible to cut off thecurrent by means of transistors M_(N12) and M_(N22) on the V_(SS) sidehaving high threshold voltages.

Embodiments 12, 13

In the embodiments shown up to this point, it has been possible toreduce the subthreshold current whether the input signal is at a lowlevel or at a high level. In a practical LSI, however, the level of aspecific signal in the period where the subthreshold current reductionis required, e.g., in a standby state is often known in advance. In suchcases, it is possible to reduce the subthreshold current by a simplercircuit.

FIG. 18 shows a circuit example of an inverter chain in which it isfound that the input signal IN in a standby state is at a low level "L".Since IN is at a low level, nodes N₁, N₃, N₅, . . . show a high level,and nodes N₂, N₄, N₆, . . . show a low level. Thus, M_(P2), M_(P4), . .. among P-channel MOS transistors are in an off state, and M_(N1),M_(N3), . . . among N-channel MOS transistors are in an off state. It issufficient to insert switches and resistances in the sources of thesetransistors in an off state because the subthreshold current flows inthe transistor in the off state.

Further, there is no problem if the switch and the resistance are heldin common by means of a plurality of inverters as shown in FIG. 19.

Although these embodiments are restricted by the fact that the level ofthe input signal has to be known, there is such an advantage that thesubthreshold current can be reduced by a simple circuit. As it becomesapparent when FIGS. 18 and 19 are compared with FIG. 11A, the number ofswitches and resistances is reduced and the level conversion circuitbecomes unnecessary.

Embodiments 14, 15

In not only an inverter, but also in a logic gate such as a NAND gateand a NOR gate, it is possible to reduce the subthreshold current by asimpler circuit when the level of the input signal in a standby statehas been known.

FIG. 20 shows an example of a two-input NAND gate, and FIG. 21 shows anexample of a two-input NOR gate. In the case when both input signals IN₁and IN₂ are at a low level or when both are at a high level, these gatesare substantially equivalent to the inverter. Consequently, the methoddescribed with reference to FIG. 18 and FIG. 19 is applicable. Theproblem exists in a case that one input is at a low level "L" and theother input is at a high level "H" as shown in the figures.

In the case of the NAND gate shown in FIG. 20, a P-channel MOStransistor M_(P12) and an N-channel MOS transistor M_(N11), are in anoff state. Since the output OUT is at a high level, however, it isM_(N11) that the subthreshold current flows in. Thus, it is sufficientto insert a switch and a resistance on the V_(SS) side. Conversely, inthe case of a NOR gate shown in FIG. 21, it is a P-channel MOStransistor M_(P14) that the subthreshold current flows in. Thus, it issufficient to insert a switch and a resistance on the V_(CC) side.

FIG. 20 and FIG. 21 show examples in which the present invention isapplied to two-input logic gates, but the present invention is alsoapplicable in a similar manner to a logic gate having three inputs ormore. Further, it is a matter of course that the switch and theresistance may be held in common with other logic gates.

Embodiment 16

FIG. 22 shows a circuit example in case it is comprehended that a clockCLK₁ is at a low level and a clock CLK₂ is at a high level in a standbystate in a clocked inverter. Since both MOS transistors M_(P16) andM_(N16) are in an off state in this case, the output OUT shows a highimpedance, and the voltage level thereof is determined by anothercircuit (not illustrated) connected to OUT. Since it is determined bythe voltage level in which of the transistors M_(P16) or M_(N16) thesubthreshold current flows, it is sufficient to insert switches andresistances on both of the V_(CC) side and the V_(SS) side in this case.

Embodiment 17

In the case of a general combinational logic circuit, it is possible toreduce the subthreshold current by a simpler circuit when the level ofthe input signal has been comprehended in advance. Description will bemade by taking the combinational logic circuit shown in FIG. 23 as anexample.

FIG. 23 shows a circuit structure example for the case where all ofinputs IN₁ to IN₆ of this circuit are at a low level. As to inverters L₁to L₃, L₅ and L₆, switches and resistances are inserted on the V_(SS)side of L₁ to L₃ and the V_(CC) side of L₅ and L₆ similarly to FIG. 18and FIG. 19. Since the input signals are all at a low level, a NOR gateL₇ is substantially equivalent to an inverter. Consequently, it issufficient to insert a switch and a resistance on the V_(SS) side. Sinceone of input signals is at a low level and the other is at a high levelwith respect to a NOR gate L₄, a switch and a resistance are inserted onthe V_(CC) side similarly to FIG. 21. Since all of three input signalsare at a low level only for L₁₂ among eight NAND gates and L₁₂ isequivalent to an inverter, a switch and a resistance are inserted on theV_(CC) side. Since input signals at a low level and at a high level areincluded for other NAND gates, it is sufficient to insert a switch and aresistance on the V_(SS) side similarly to FIG. 20.

As is apparent from the above description, it is sufficient to insert aswitch and a resistance on the V_(SS) side for a logic gate having anoutput at a high level and on the V_(CC) side for a logic gate having anoutput at a low level. The layout area can be saved by holding theseswitches and resistances in common by a plurality of logic gates.

Embodiment 18

It is also possible to reduce the subthreshold current by a simplercircuit as for a circuit with feedback in case the level of a signal isknown in advance. FIG. 24 shows an example in which the presentinvention is applied to a latch circuit shown in FIG. 16A.

In a latch circuit of this sort, both input signals IN₁ and IN₂ arenormally at a low level in a standby state, and one of output signalsOUT₁ and OUT₂ is brought to a high level and the other is brought to ahigh level, thus holding information in one bit. FIG. 24 shows a circuitstructure example in case it is comprehended that OUT₁ is at a low leveland OUT₂ is at a high level. A NAND gate L₁ is equivalent to an invertersince two input signals thereof are both at a high level, and a switchand a resistance are inserted on the V_(CC) side similarly to FIG. 18and FIG. 19. It is sufficient to insert a switch and a resistance on theV_(SS) side similarly to FIG. 20 for a NAND gate L₂ since one of inputsignals thereof is at a low level and the other is at a high level. Itis a matter of course that these switches and resistances may be held incommon with other logic gates.

Embodiment 19

FIG. 25 shows an example in which the present invention is applied to awell known data output buffer in a memory LSI or the like. In a standbystate, an output enable signal OE is at a low level, outputs of NANDgates L₂₁ and L₂₂ are at a high level and an output of an inverter L₂₃is at a low level. Accordingly, two MOS transistors M_(P20) and M_(N20)constituting an output stage L₂₄ are both in an off state, and an outputDOUT has a high impedance.

As to logic gates L₂₁ to L₂₃, it is sufficient to insert a switch and aresistance on the V_(SS) side or the V_(CC) side in accordance with thepolicy stated in the description with reference to FIG. 23. As to anoutput stage L₂₄, it is sufficient to insert switches and resistances onboth the V_(CC) side and the V_(SS) side in a similar manner to the caseof the clocked inverter shown in FIG. 22.

Embodiment 20

FIG. 26 shows an example in which the present invention is applied to awell known data input buffer in a memory LSI or the like. In FIG. 26, SBrepresents a signal which shows a high level in a standby state. Outputsof inverters L₃₁ and L₃₂ can be used as φ_(S) and φ_(C) for controllingswitches respectively as shown in FIG. 4 and FIG. 7. L₃₃ represents aNAND gate and receives φ_(S) and a data input signal D_(IN). Since φ_(S)is at a low level in a standby state, the output of L₃₃ shows a highlevel irrespective of D_(IN). Thus, an output d_(IN) of an inverter L₃₄shows a low level. On the other hand, since SB is at a low level in anoperating state, d_(IN) follows in the wake of D_(IN).

Concerning the NAND gates L₃₃ and the inverter L₃₄, the subthresholdcurrent can be reduced by inserting switches and resistances on theV_(SS) side and the V_(CC) side, respectively. Although such techniquescannot be applied to the inverters L₃₁ and L₃₂, the subthreshold currentcan be reduced by enhancing the threshold voltages of the MOStransistors. Since high-speed performance is not required in many casesfor changing over the standby state to and from the operating state,there is no problem in using MOS transistors having enhanced thresholdvoltages.

A data input buffer has been described above, but the same is applied toan input buffer for an address signal and other signals.

The embodiments illustrated in FIGS. 18 to 25 have a merit that thesubthreshold current can be reduced by a simple circuit, but on theother hand, these embodiments are restricted by that they areunapplicable unless the signal level in a period where subthresholdcurrent reduction is required, e.g., in a standby state, is known.Accordingly, it is desirable at this time to settle the levels of asmany nodes as possible in the LSI. It is possible to have the level ofthe signal d_(IN) at this time settled to a low level by using a circuitsuch as the input buffer shown in FIG. 26 as means for the above. Asanother method for deciding upon the level, there is also a methodwherein the data input terminal D_(IN) is specified to be a low level(or a high level) in case of a standby state.

The embodiments illustrated in FIG. 18 to FIG. 26 are suitable forapplication to a memory LSI. Because, in the memory LSI, there arecomparatively many nodes in which either a high level or a low level isknown at time of standby state, and the levels of most nodes can besettled by using the input buffer shown in FIG. 26.

In the random logic LSI such as a microprocessor, it is effective to fixthe voltage of a troublesome node forcibly by fixing the output of aninternal register or by adding a logic such as a flip-flop having aresetting function. FIG. 35 shows an embodiment of the structure of thelatch capable of fixing an output. This circuit is simplified byreplacing the inverter in the ordinary latch by a NAND circuit. Asillustrated in FIG. 36, the latch operates as an ordinary one while thesignal φ_(S) is at the high level, and the level of the output signal Qis fixed to the high level while the signal φ_(S) is at the low level(or in a sleep mode). Here, the sleep mode is one for interrupting theoperation of the entire LSI or the circuit block unit so as to reducethe current dissipation. Incidentally, the subthreshold current of thelatch itself can be reduced in the sleep mode if the signal φ_(t) is atthe low level whereas the signal φ_(b) is at the high level. If thislatch is used, the node N₄₁ is forcibly set to the high level becausethe signal φ_(S) takes the low level, so that the data are erased fromthe register in the sleep mode. However, this erasure raises no problemeven in the use, in which the necessary data in the CPU are saved to themain memory to open the reset state again after the sleep mode, that is,for the resume function in which a notebook personal computer is held inthe standby state if it receives no input for a predetermined period.FIG. 37 shows another embodiment of the latch capable of fixing theoutput forcibly. As shown in FIG. 38, this circuit also acts as anordinary latch while the signal φ_(S) is at the high level and fixes thelevel of the output signal Q to the high level while the signal φ_(S) isat the low level. This latch can retain the data even in the sleep modebecause the node N₄₁ is not influenced even if the signal φ_(S) takesthe low level. The operation can be reopened from the state before thesleep mode after this sleep mode is released and can establish the sleepmode even while the CPU is executing its task. Thus, this embodiment issuitable for the case in which the operation is resumed after arelatively short time from the sleep mode.

The embodiments illustrated in FIGS. 25 and 26 can be used not only asan input-output circuit for an external terminal of an LSI chip, butalso as driver/receiver for an internal bus of a microprocessor forinstance.

Embodiment 21

The embodiments in which the present invention is applied to a CMOScircuit have been described so far, but the present invention is alsoapplicable to a circuit composed of MOS transistors having a singlepolarity. FIG. 27 shows an example of a circuit composed of N-channelMOS transistors only. In FIG. 27, PC represents a precharge signal, andIN₁ and IN₂ represent input signals.

At the time of standby, i.e., in a precharge state, PC is at a highlevel and IN₁ and IN₂ are at a low level, and the output OUT isprecharged to a high level (=V_(CC) -V_(T)). At the time of operation,IN₁ and IN₂ are brought to a high level or remain at a low level afterPC is brought to a low level. When at least one of IN₁ and IN₂ isbrought to a high level, OUT is brought to a low level. When both of IN₁and IN₂ remain at a low level, OUT is left (as is) at a high level.Namely, this circuit outputs the NOR of IN₁ and IN₂.

In this circuit, M_(N41) and M_(N42) on the V_(SS) side are thosetransistors that are in an off state at time of standby, and thesubthreshold current flows in these transistors. Accordingly, in orderto apply the present invention to this circuit, it is sufficient toinsert a switch and a resistance on the V_(SS) side as shown in thefigure. They are not required on the V_(CC) side.

Incidentally, in the LSI for complicated operations such as a randomlogic LSI, the logic (or voltage) state of each node in the chip in thestandby state, for example, is determined by the design automation (DA)method so that the position to insert the aforementioned switch andresistor can be automatically determined by the DA.

As described above, the present invention is very effective forachieving low power consumption of MOS transistor circuits and asemiconductor integrated circuit composed of the same. The demand forlow power consumption of a semiconductor integrated circuit is great,and recently a microprocessor system having a low power backup mode wasdescribed in the Sep. 2, 1991, issue of Nikkei Electronics, pp. 106-111,for instance. In the backup mode, the clock is stopped and the supply ofpower to unnecessary parts thereof is suspended, thereby reducing powerconsumption. However, no consideration is given to the extent ofreduction of the subthreshold current. These processor systems operateat 3.3 to 5 V and can use transistors having a sufficiently highthreshold voltage so that the subthreshold current to too low to raiseany problem. However, if the operating voltage becomes as low as 2 or1.5 V so that the threshold voltage has to be dropped, the excessivesubthreshold current cannot be reduced any more by the method of theprior art using the CMOS circuit. When the present invention is appliedto, for example, a resuming circuit to which the power is supplied evenin the backup mode, lower power consumption can be realized.

Embodiment 22

In the examples described above, there are such problems that the logicvoltage swing is reduced with the increase of the number of stages, anda more or less complicated design is required for the case where thevoltage level of an input signal is unknown. FIG. 28 shows a circuit forsolving these problems, in which the switches are switched on so as toperform normal high-speed operation in a period required until the logicoutput is settled as described so far. In other periods than the above,a subthreshold current passage of a logic circuit is cut off byswitching off the switches. However, since a supply passage of the powersupply voltage is interrupted when the switches are switched off, theoutput of the logic circuit becomes floating, and the logic output is nolonger settled. Thus, it is a feature that a sort of latch circuit (alevel-hold circuit) for holding a voltage level is provided at theoutput thereof. If a transistor having a high threshold voltage or thelike is used for the level-hold circuit, the subthreshold current of thelevel-hold circuit becomes negligibly small, thus making it possible tomake the subthreshold current small on the whole. The delay time isaffected little by the level-hold circuit, and is determined by thelogic circuit. Even if a high-speed circuit having large drivingcapability is used in the logic circuit, the consuming current is onlythe current flowing through the level-hold circuit since no currentflows through the logic circuit in a standby state. The level-holdcircuit may have a small driving capability since it only holds theoutput, thus making it possible to reduce the current consumption. Sincethe output of the logic circuit is held by the level-hold circuit evenif the switches are switched off, there is no possibility of inversionof the output and the operation is stabilized. Thus, a semiconductordevice operating stably with low power consumption and at a high speedcan be realized. According to the present embodiment, since the voltagelevel is always guaranteed at a constant value by means of thelevel-hold circuit, the logic voltage swing will never be decreased withthe increase of the number of the logic stages. Further, the presentembodiment is effective independent of the logic input.

The present embodiment will be described further with reference to FIG.28. A logic circuit LC is connected to a power supply line VHH at a highpotential and a power supply line VLL at a low potential throughswitches SWH and SWL. Here, it is also possible to have VHH and VLLcorrespond to V_(CC) and V_(SS) which have been heretofore described,respectively. A level-hold circuit LH is connected to an output terminalOUT of the logic circuit LC. Switches SWH and SWL are controlled by acontrol pulse CK so as to be switched on and off at the same time. Thelogic circuit LC is composed of a logic gate such as an inverter, a NANDcircuit, a NOR circuit, a flip-flop circuit, or a plurality ofcombinations thereof. The level-hold circuit LH can be composed of apositive feedback circuit.

The operation of the logic circuit LC is performed with the switches SWHand SWL on. After an output OUT in accordance with an input IN of thelogic circuit LC is settled or determined, the switches SWH and SWL areswitched off, a current passage from VHH to V_(SS) through the logiccircuit LC is cut off, and the output of the logic circuit LC is held bythe level-hold circuit LH.

The delay time of a circuit is affected little by the level-hold circuitLH, and is determined by the logic circuit LC. It is possible to performa high-speed operation having a short delay time by using a circuithaving large driving capability for the logic circuit LC. For example,in a standby state, since no current flows through the logic circuit LC,the consuming current is the only one that flows through the level-holdcircuit LH. Since a level-hold circuit LH having small drivingcapability will suffice, the consuming current can be made small.Moreover, since the output OUT of the logic circuit LC is maintained bymeans of the level-hold circuit LH, there is no possibility ofmalfunction. Hence, a circuit which performs stabilized operation withlow power consumption and at a high speed can be realized.

Embodiment 23

An embodiment in which the present invention is applied to a CMOSinverter is shown in FIG. 29. An NMOS transistor MN1 and a PMOStransistor MP1 operate as the switches SWL and SWH shown in FIG. 28,respectively. In order to reduce a leakage current when the switches areswitched off, the threshold voltages of the transistors MN1 and MP1 aremade sufficiently high. Their channel width/length ratios are determinedso that on state resistances do not become large. A control pulse CK isinputted to the gate of the NMOS transistor MN1 and a control pulse CKBis inputted to the date of the PMOS transistor MP1. CKB represents acomplementary signal of CK. A CMOS inverter INV composed of an NMOStransistor MN2 and a PMOS transistor MP2 is connected to MN1 and MP1. Inorder to increase the driving capability in low voltage operation, thethreshold voltages of the transistors MN2 and MP2 are made low. Thelevel-hold circuit LH composed of NMOS transistors MN3 and MN4 and PMOStransistors MP3 and MP4 is connected to an output terminal OUT of theinverter INV. In order to reduce a penetrating current while holding theoutput, the threshold voltages of the transistors MN3, MN4, MP3 and MP4are made sufficiently high, and the channel width/length ratios thereofare made sufficiently small. An example of numeric values of the powersupply voltage and the threshold voltages will be shown. VLL is set tothe ground potential at 0 V, and VHH is set to the external power supplyvoltage at 1 V. The threshold voltages of the NMOS transistors are setto 0.2 V for MN2 and to 0.4 V for MN1, MN3 and MN4. The thresholdvoltages of the PMOS transistors are set to -0.2 V for MP2 and to -0.4 Vfor MP1, MP3 and MP4.

The operation will be described with reference to a timing chart shownin FIG. 30. First, the control pulse CK is raised to VHH, CKB is loweredto VLL, transistors MN1 and MP1 are turned on, and the inverter INV isconnected to VHH and VLL. When the input signal IN is raised to VHH fromVLL, MP2 is turned off and MN2 is turned on, and the output OUT isdischarged from VHH to VLL. The transistor MN2 starts conduction in asaturated region, and the value of the current flowing in MN2 isdetermined by the voltage between the gate (input terminal IN) and thesource (node NL). Since the transistor MN1 is provided between the nodeNL and VLL, the potential of the node NL rises temporarily by theon-state resistance of MN1 and the current flowing from MN2. Since thegate of MN1 is at VHH, however, a design is possible so that theon-state resistance becomes sufficiently small even if the thresholdvoltage is high, thus reducing the influence upon the delay time.Further, MN4 is in an off state and MP4 is in an on state so that thelevel-hold circuit LH holds the output OUT at VHH when the output OUT isinverted to VLL. As a result, a current flows from VHH to VLL throughMP4 and MN2 since MN2 is turned on, but the influence exerted on thedelay time and the current consumption is small by designing the drivingcapability of MP4 small as compared with that of MN2. When the outputOUT lowers, MN3 is turned off and MP3 is turned on, a node NLH in thelevel-hold circuit is inverted from VLL to VHH, MN4 is turned on and MP4is turned off, and the level-hold circuit LH operates so as to hold theoutput OUT at VLL, thus preventing current from flowing. MP2 is in anoff state since the gate and the source are both at VHH, but the leakagecurrent is large and current flows through the inverter INV since thethreshold voltage is low. Then, the control pulse CK is lowered to VLL,CKB is raised to VHH, and the transistors MN1 and MP1 are turned off,thus isolating the inverter INV from VHH and VLL. At this time, MN1 andMP1 are turned off completely because the gate and the source thereofare at equal potential and the threshold voltage is high. The output OUTis held at VHH by positive feedback of the level-hold circuit LH. Sincethe NMOS transistor MN2 is in an on state, the node NL is held at VLL.On the other hand, the voltage of a node NH starts to drop due to theleakage current of the PMOS transistor MP2 from the node NH to theoutput terminal OUT. Then, the source potential drops lower than thegate potential, and MP2 is turned off completely. As a result, thecurrent of the inverter INV does not flow in a standby state. Further,before the input signal IN changes, the control pulse CK is raised toVHH, CKB is lowered to VLL, and the transistors MN1 and MP1 are turnedon, thus bringing the node NH to VHH. Since the input IN is invertedfrom VHH to VLL, the output OUT is inverted from VLL to VHH.

It is desirable that the level-hold circuit LH follows the output OUTquickly so that the period of time when the current flows through theinverter INV and the level-hold circuit LH becomes shorter.Consequently, the inverter INV and the level-hold circuit LH arearranged close to each other so as to reduce the wiring delay.

As is apparent from the present embodiment, when the threshold voltageof the MOS transistor used as the switch is brought to approximately 0.4V or higher which has been deemed to be required to reduce thesubthreshold current, it is possible to lower the threshold voltage ofthe MOS transistor in the logic circuit without increasing the currentflow in a standby state. Even if the operating voltage is lowered to 1 Vor lower, it is possible to set the threshold voltage of the MOStransistor to 0.25 V or below, thereby to secure driving capability.Thus, low power consumption by lowering of voltage can be realized.Further, it is possible to realize performance improvement by thescaling down of elements based on a conventional scaling law. Moreover,since the structure is the same as a conventional CMOS logic circuitexcept that the switch and the level-hold circuit are loaded, the samedesign techniques as before can be used.

Embodiment 24

FIG. 31 shows an embodiment in which the present invention is applied toa CMOS inverter chain. An inverter chain can be realized by a multistageconnection of two switches and also a level-hold circuit provided on theinverter at one stage as shown in FIG. 29. Also, by the presentembodiment, the switches and level-hold circuit are held in common by aplurality of inverters so as to reduce the number of elements and thearea. A case of an inverter chain with four stages is taken as anexample here, but a case including a different number of stages is alsostructured in a similar manner. Four inverters INV1, INV2, INV3 and INV4are cascaded. The level-hold circuit LH is connected to the outputterminal OUT of the inverter INV4 at the last stage. Each inverter iscomposed of one PMOS transistor and NMOS transistor similarly to the INVshown in FIG. 29. The transistor size in respective inverters may eitherbe the same as or different from each other. With a cascaded driver, itis also possible to increase the channel width in the order of INV1,INV2, INV3 and INV4 at a constant interstage ratio while keeping thechannel length the same. The sources of the PMOS transistors ofrespective inverters are connected to the node NH, and the sources ofthe NMOS transistors thereof are connected to the node NL. A switch SWLis provided between the node NL and the power supply VLL on a low level,and a switch SWH is provided between the node NH and the power supplyVHH on a high level. The switches SWL and SWH are controlled by acontrol pulse CK and switched on and off at the same time. As shown inFIG. 29, the switch SWL is realized by an NMOS transistor, and theswitch SWH is realized by a PMOS transistor applied with a complementarysignal of CK at the gate thereof.

The operation of the inverter chain is performed with the switches SWLand SWH on. For example, when the output IN is inverted from the lowlevel VLL to the high level VHH, a node N1 is inverted from VHH to VLLby means of the inverter INV1, a node N2 is inverted from VLL to VHH bymeans of the inverter INV2, a node N3 is inverted from VHH to VLL bymeans of INV3, and the output OUT is inverted from VLL to VHH by meansof INV4. When OUT is at VHH, the level-hold circuit LH is operated so asto hold OUT at VHH. In a standby state, the current passage from VHH toVLL through the inverters is cut off by switching off the switches SWLand SWH.

It is sufficient to provide a level-hold circuit at the output terminalthereof only by handling the inverter chain collectively as one logiccircuit as in the present embodiment. Further, the switches SWL and SWHcan be held in common by a plurality of inverters. The sizes of theswitches SWL and SWH are determined by the magnitude of the peak currentapplied. The peak of the current sum flowing in a plurality of invertersbecomes smaller than the sum of peak currents in respective inverters.For example, when an inverter chain is formed assuming an interstageratio at 3, the peak of the current sum becomes almost equal to the peakcurrent at the last stage. Accordingly, a small area of the switch willsuffice when the switch is held in common by a plurality of inverters ascompared to a case where a switch is provided for every inverter.

Embodiment 25

FIG. 32 shows another embodiment in which the present invention isapplied to an inverter chain. A case of an inverter chain at four stagesis taken as an example similarly to FIG. 31, but the inverter chain isstructured similarly in the case of another number of stages, too.Inverters INV1, INV2, INV3 and INV4 are connected in series. Level-holdcircuits LH3 and LH4 are connected to a node N3 which is an outputterminal of the inverter INV3 and an input terminal of the inverter INV4and to an output terminal OUT of INV4, respectively. Each inverter iscomposed of one each of a PMOS transistor and an NMOS transistorsimilarly to the INV shown in FIG. 29. Odd-numbered inverters INV1 andINV3 are connected to nodes NL1 and NH1, and even-numbered invertersINV2 and INV4 are connected to nodes NL2 and NH2. Switches SWL1 and SWL2are provided between the nodes NL1, NL2 and the power supply VLL on alow level side, respectively, and switches SWH1 and SWH2 are providedbetween the nodes NH1, NH2 and the power supply VHH on a high levelside, respectively. The switches SWL1, SWL2 and SWH1, SWH2 arecontrolled by a control pulse CK, and switched on and off at the sametime.

The operation of the inverters is performed with the switches SWL1,SWL2, SWH1 and SWH2 on. For example, when the input IN is inverted fromthe low level VLL to the high level VHH, a node N1 is inverted from VHHto VLL, a node N2 is inverted from VLL to VHH, a node N3 is invertedfrom VHH to VLL, and the output terminal OUT is inverted from VLL to VHHby means of INV4 in consecutive order. When N3 is at VLL, the level-holdcircuit LH3 operates so as to hold N3 at VLL. Further, when OUT is atVHH, the level-hold circuit LH operates so as to hold OUT at VHH. Forexample, in a standby state, the current passages from VHH to VLLthrough the inverters are cut off by switching off the switches SWL1,SWL2, SWH1 and SWH2. At this time, since the node N3 is held at a lowlevel VLL by the level-hold circuit LH3, the node NL1 is also held atVLL through the inverter INV3. Furthermore, the node N1 is held at VLLthrough the inverter INV1. Similarly, nodes NH2 and N2 are also held atVHH because the output terminal OUT is held at the high level VHH bymeans of the level-hold circuit LH4. Thus, the nodes connecting theinverters are held at either VHH or VLL.

As described above, the nodes N1, N2 and N3 among the inverters are allheld at either a high level or a low level by providing two sets ofswitches, connecting odd-numbered inverters and even-numbered invertersto different switches and connecting the level-hold circuits to anyoutput terminal of odd-numbered inverters and any output terminal ofeven-numbered inverters, respectively. Since the input to each inverteris not brought to an intermediate level even if the standby statecontinues for a long time, operation is stabilized, and neither datainversion nor DC current flow does not occur when the switches areturned on.

In the embodiments shown in FIGS. 29-32, the present invention isapplied to a CMOS inverter and an inverter chain. However, the presentinvention is not limited to these embodiments described so far, but alsoincludes other embodiments not departing from the gist that stabilizedoperation is performed with low power consumption and at a high speed byloading a logic circuit with a switch and a level-hold circuit.

Embodiment 26

For example, another embodiment in which the present invention isapplied to a CMOS inverter is shown in FIG. 33. In the embodiment shownin FIG. 29, transistors MN1 and MP1 operating as switches are providedbetween the CMOS inverter INV and power supplies VLL and VHH. Incontrast to the above, these transistors are provided between an NMOStransistor and a PMOS transistor in the present embodiment.

Two NMOS transistors MN2 and MN1 and two PMOS transistors MP1 and MP2are connected in series between the low level power supply VLL and thehigh level power supply VHH. The NMOS transistor MN1 and the PMOStransistor MP1 operate as switches. In order to reduce the leak currentwhen those transistors are turned off, the threshold voltages of thetransistors MN1 and MP1 are made high. A control pulse CK is inputted tothe gate of the NMOS transistor MN1 and a control pulse CKB of acomplementary signal of CK is inputted to the gate of the PMOStransistor MP1. The NMOS transistor MN2 and the PMOS transistor MP2 areconnected to the input terminal IN at the gates thereof, and operate asCMOS inverter. In order to increase the driving capability in lowvoltage operation, the threshold voltages of the transistors MN1 and MP1are made low. A level-hold circuit LH structured similarly to FIG. 29 isconnected to the output terminal OUT.

The operation is performed in a similar manner as the embodiment shownin FIG. 29. The transistors MN1 and MP1 are turned on by control pulsesCK and CKB, thus having MN2 and MP2 operate as a CMOS inverter. Forexample, when the input IN is inverted from the low level VLL to thehigh level VHH, the transistor MN2 which has been in an off state startsto conduct and operates in a saturated region. At this time, the currentvalue of MN2 is determined by the gate-source voltage. Since thetransistor MN1 is provided between MN2 and the output terminal OUT, theon-state resistance of MN1 is connected to the drain of MN2. As aresult, the influence of the on-state resistance of MN1 exerted upon thecurrent value of MN2 is small. After the output OUT is determined, thetransistors MN1 and MP1 are turned off, thereby to prevent furthercurrent flow and maintain the output OUT by means of the level-holdcircuit LH.

When switches are inserted on the output terminal side of the logiccircuit as in the present embodiment, it is impossible to hold theswitches in common by means of a plurality of logic gates, but theinfluence of the on-state resistance of the switches is small. In casethe transistors used as switches are the same, the delay time is reducedas compared with a case where the switches are provided on the powersupply side of the logic circuit as in the embodiment shown in FIG. 29.Otherwise, if it is designed so that the delay time becomes equal, thechannel width/length ratio of the transistor used as a switch may bereduced, thus enabling a reduction in area.

Embodiment 27

FIG. 34 shows another structure example of the level-hold circuit. Acase in which this level-hold circuit is replaced with a level-holdcircuit composed of NMOS transistors MN3 and MN4 and PMOS transistorsMP3 and MP4 in the embodiment shown in FIG. 29 will be described.

This level-hold circuit is composed of NMOS transistors MN3, MN4 and MN5and PMOS transistors MP3, MP4 and MP5. In order to reduce the leakagecurrent in a standby state, the threshold voltages of respectivetransistors are made high. For example, it is set to 0.4 V for the NMOStransistor and -0.4 V for the PMOS transistor. MN3 and MP3 form asinverter, and MN4, MN5, MP4 and MP5 form a clocked inverter. A controlpulse CKB is inputted to the gate of MN5, and a control pulse CK isinputted to the gate of MP5. The operation timing is the same as thecase when the level-hold circuit LH shown in FIG. 29 is used as shown inFIG. 30. The control pulse CK is raised to the high level VHH and CKB islowered to the low level VLL so as to operate the inverter INV. At thistime, the transistors MN5 and MP5 are turned off in the level-holdcircuit. As a result, when the output OUT is inverted, current does notflow through the inverter INV and the level-hold circuit, and delay timeand current consumption may be reduced. In a standby state, the controlpulse CK is lowered to the low level VLL and CKB is raised to the highlevel VHH, and the inverter INV is isolated from power supplies VLL andVHH. At this time, the transistors MN5 and MP5 are turned on in thelevel-hold circuit, and the output OUT is held by positive feedback.

By forming the level-hold circuit by the combination of an inverter anda clocked inverter as described above, the number of transistors isincreased by two, but the logic circuit and the level-hold circuit do nolonger conflict with each other, and the delay time and the currentconsumption may be reduced. Further, the driving capability of thelevel-hold circuit may also be increased, and there is no possibilitythat the output fluctuates even when the leakage current through theoutput terminal is large, thus making stabilized operation possible.

FIG. 39 shows an embodiment in which this invention is applied to alogic circuit for logic operations with a dual-phase clock. In the LSIof an ordinary microprocessor, most of the logic operations in the chipare frequently synchronized with a dual-phase clock. The logic circuitis divided into the logic circuits LC1 and LC2 which are additionallyequipped at their individual outputs with latches LT1 and LT2 to becontrolled by clocks CK1b and CK2b. In this embodiment, the latches LT1and LT2 play the role of the level holder. Here, the logic circuits LC1and LC2 are combinational logic circuits each of which is composed ofone or a plurality of logic gates. These two logic circuits LC1 and LC2are alternately operated in synchronism with the clocks so that switchesSWH1 and SWL1 and switches SWH2 and SWL2 are alternately turned ON andOFF by the clocks to interrupt the subthreshold current of the logiccircuit which is not operated. According to this embodiment, it ispossible to realize a low-power LSI having a low operating voltage and asmall subthreshold current.

The operations will be described by using a specific circuit embodimentshown in FIG. 40 and the timings of the control clocks illustrated inFIG. 41. For simplicity, the logic circuits LC1 and LC2 are individuallyexemplified by one inverter. Moreover, the latches LT1 and LT2 areexemplified by the level holder shown in FIG. 34 but may be exemplifiedby the circuit shown in FIG. 29. Clocks CK1t and CK2t alternately takethe high level without any mutual super-position. The clocks CK1b andCK2b are the signals which are inverted from the clocks CK1t and CK2t,respectively. Here, the high-speed operations can be achieved if the MOStransistors composing the logic circuits LC1 and LC2 have theirthreshold voltages set at the low level. On the other hand, the MOStransistors having their gates supplied with the clocks must be able tointerrupt the subthreshold current when turned OFF. For this necessity,it is sufficient to raise the threshold voltages or to set the highlevel of the clocks higher than the high voltage VHH and the low levellower than the low voltage VLL.

In the operation mode, the logic circuit LC1 is operated while the clockCK1t is at the high level. At this time, the clock CK2t is at the lowlevel so that the latch LT2 latches the data to be supplied thereto.Moreover, since the logic circuit LC2 need not operate, the transistorsMP12 and MN12 are turned OFF to interrupt the subthreshold current.While the clock CK2t is at the high level, on the contrary, the latchLT1 latches the data, and the logic circuit LC2 operates so that thesubthreshold current of the logic circuit LC1 can be interrupted. Inshort, the current of either the logic circuit LC1 or LC2 1can beinterrupted to reduce the subthreshold current to one half of the priorart.

In order to reduce the power in the recent microprocessor operating at3.3 V to 5 V, as described above, the charging and discharging currentsare reduced by interrupting the supply of clocks to an unnecessarycircuit in the low-power backup mode (i.e., the sleep mode). In thisembodiment, when both the clocks CK1t and CK2t are held at the low levelin the sleep mode, as shown in FIG. 41, both the transistors MP11 andMN11 and the transistors MP12 and MN12 are turned OFF to interrupt thecurrent flow through both the logic circuits LC1 and LC2. As a result,the effect for reducing the subthreshold current in the sleep mode isfar higher than that in the operation mode.

FIG. 42 shows another embodiment of this invention, in which theinvention is applied to a gate array. This gate array is a digital logiccircuit so that the foregoing embodiments can be applied to reduce thesubthreshold current. In the gate array, however, some gates are leftunused and inactivated when the logic circuit is to be constructed, aswill be described in the following. FIG. 42(a) shows the embodiment inwhich the logic shown in FIG. 42(b) is constructed in a one-circuitblock of the gate array using two-input NANDs as its basic cells. Inthis Figure, blocks A001, A002 and A003 enclosed by broken lines are thebasic NAND cells. Moreover, INN1 and OUT1 designate the input and outputof this logic circuit block, respectively. In case the inverter isconstructed of the NAND cells, as shown, it is a current practice to fixinputs A004 and A005 at the high level (V_(CC)) thereby to inactivatethe corresponding gates. The gates thus inactivated frequently occupyseveral ten % of the usable gates. Thus, the subthreshold currents thatflow through the inactivated gates cannot be ignored in the low-voltagegate array in which the threshold voltages of the transistors are scaleddown. As shown, transistors MA01 and MA03 have their sources connectedwith a second power source line V_(CL) which is separated from the powersource V_(CC) through the transistor M_(C) and the resistor R_(C). Inthe low-power mode, the signal φ_(C) is set to the high level to cut offthe transistor M_(C). Then, the transistors MA01 and MA03 are inverselybiased between their gates and sources and are deeply cut off so thatthe subthreshold currents of the inactivated gates can be drasticallyreduced. As to the active gates, however, the leakage currents can alsobe prevented, if the p-channel transistors have their sources connectedwith V_(CC) or V_(CL) whereas the n-channel transistors have theirsources connected with V_(SL) or V_(SS) in accordance with the logicstates (at the high level "H" or the low level "L", as shown) of theindividual gate outputs for a period requiring the low lower dissipationas in the standby state. Incidentally, as to the inactive gate, nocurrent has to be supplied to the transistors even when they are inoperation so that an interconnection other than the power source lineV_(CL) formed to have a minimum interconnection width and a highimpedance can be used. For this interconnection, the transistor M_(C) isnot always essential, and only use of the resistor R_(C) may besufficient.

FIGS. 43A and 43B show another embodiment of this invention, in whichthe prevention of the subthreshold current according to this inventionis applied to the inactive gates in a gate array composed of basic cellsof two-input NORs. FIG. 43A shows the embodiment in which the logicshown in FIG. 43B is composed of the NOR cells. In FIG. 43A, brokenlines A001, A012 and A013 are the basic NOR cells. In case the inverteris composed of the NOR cells, it is a current practice to fix an inputA014 or A015 at the LOW (V_(SS)) level thereby to inactivate thecorresponding gates. If the transistors MA11 and MA13 have their sourcesconnected at this time with the lower source line V_(SL), they can bedeeply cut off on the basis of the aforementioned operating principle,to prevent the subthreshold current.

Moreover, as the LSI chip has its scale enlarged, a test circuit fortesting other circuit groups is packaged on the chip. This test circuitcan have its operation interrupted during ordinary chip operation (otherthan the testing time). In this case, the foregoing embodiments areeffective for reducing the subthreshold current of the test circuit.

An example in which the individual embodiments thus far described areapplied to a single-chip microprocessor is described next. First of all,a microprocessor having the foregoing power reducing mechanisms will bedescribed. In the microprocessor of the prior art, the power iscontrolled by controlling the entire chip at one time. Since the chipi386SL of Intel, for example, has a completely static internal circuit,its internal status can be retained, even if the input of a clock to thechip is interrupted, to reopen the operation when the input of the clockis applied again. Thus, the operation of the entire chip is interruptedto reduce the power of the entire system by interrupting the input ofthe clock. However, this is possible only when the power source voltageis as high as 3.3 V to 5 V as in the prior art. This is because the MOStransistors composing the CMOS circuit can take threshold voltages ashigh as about 0.4 to 0.5 V so that their subthreshold currents can bereduced to negligible values. As has already been describedthereinbefore, however, the high-speed system to be operated by a powersource voltage of 2 V or less or by the voltage of one battery of about0.9 or 1.6 V cannot have its power reduced any more even if the clock isinterrupted. In the LSI composed of logic gates, e.g. mainly randomgates, generally speaking, the number of the logic gates having varyinginput voltages is about 20% of all the multiple logic gates in the chip.The remaining about 80% logic gates have their inputs unvaried. Sincethe CMOS circuit of the prior art advantageously has a high thresholdvoltage, most of the power of the 80% of the logic gates can beneglected to reduce the power of the entire chip. However, thisadvantage cannot be expected any more at a low power source voltage.Next will be described a microprocessor as one example of an electronicapparatus in which the entire chip can have its power reduced foroperation at a low power source voltage.

FIG. 44 shows a single-chip microprocessor having a power reducingmechanism of this invention packaged therein. This microprocessor isfeatured in that the chip is equipped with a mechanism for controllingthe active/standby modes for each unit, as will be described in thefollowing. Reference numeral 600 designates the single-chipmicroprocessor. This microprocessor 600 is packaged thereover with acentral processing unit (CPU) 601, a co-processor A (COPA) 602, aco-processor B (COPB) 603, a local memory (LM) 604 and a bus controlunit (BUSC) 605. These individual units are connected through aninternal bus 651 over the chip. Moreover, the units are connected withthe chip externally through the BUSC 605 and an external bus 652. Withthis external bus 652, there are connected a main memory (MS) 606, aninput/output device (IO) 607 and so on. The CPG 606 is a clockgenerator, and the individual units in the chip operate in synchronismwith a clock signal 653 generated by the CPG 606.

The COPA 602, the COPB 603 and the LM 604 individually have twooperating states, one of which is the sleep state. In this state, theindividual units do not operate so that the power to be dissipated isremarkably reduced. Another is the active state. In this state, theunits execute the data reading/writing operations and other executeoperations. Therefore, the power dissipation is not suppressed to aremarkably small value. The logic circuits constructed of theseindividual units are exemplified by the circuits of FIGS. 18 to 26,FIGS. 28 to 32 and FIGS. 39 to 41. As a result, the power dissipation inthe sleep state can be reduced. In the active state, too, the power canalso be reduced by controlling the active state finely for each phase ofthe dual-phase clocks with the circuits of FIGS. 39 to 41, for example.The MS 606 and the IO 607 have the active/sleep states, too. Signals 654and 655 to be outputted from the microprocessor 600 are signals forinstructing to activate the MS 606 and the IO 607, respectively.

The co-processors COPA 602 and COPB 603 are basically the units of thesame kind and execute the specified operations only when the programexecuted in the CPU contains an instruction to require the execution ofthe COPA or COPB. It is sufficient that the active state is effectedonly at that time while the sleep state prevails in the remainingperiod. In an ordinary program, the request for this execution is not sofrequent. These co-processors are characterized in that they areequipped with a number of register files and a (or a plurality of)proprietary execution unit so that they have a number of transistors intheir entirety.

On the other hand, the LM 604 is stored with the programs and datarequired by the CPU so that the frequency of its access is high. Incase, however, the CPU has a cache memory packaged therein, theexecution is performed in the CPU. This structure is characterized inthat the accessing frequency is dropped to elongate the period of thesleep state.

The CPU 601 is a unit for executing instructions and processing data andfor executing programs at all times (at an activity ratio of 100%). TheCPU contains the fundamentals of an ordinary processor, such as ageneral purpose register, an execution unit and so on and may contain acache memory, as the case may be. The instructions or data are stored inthe LM 604 or MS 606. The LM 604 is an on-chip memory having a smallcapacitance but accessible at a high speed so that it is stored with theinstructions or data to be frequently used by the CPU 601. Theinstructions or data, which need not be accessed so frequently, arestored in the memory MS 606 which has a large capacitance but anintermediate speed. The CPU 601 can access the LM 604 directly throughthe internal bus 651. On the other hand, the access to the MS 606 isexecuted through the internal bus 651, the BUSC 605 and the external bus652. The BUSC 605 includes external bus buffers having a bit width ofabout 32 to 128 bits. The BUSC may be activated only for accessing amemory or device outside of the chip. The BUSC may be held in the sleepstate if the program or data it needs are present in the chip.

The COPA 602 is a co-processor for executing multiplication, divisionand calculations of roots and absolute values and has a proprietaryexecution unit packaged therein for processing these calculations athigh speed. The COPB 603 is a co-processor for executing functionalcalculations of trigonometric functions and distances and has aproprietary execution unit for processing those calculations at a highspeed. The CPU 601 starts the executions of the individual co-processorsby writing commands for instructing the operations required into commandregisters CMDA 609 and CMDB 610 in the COPA 602 and COPB 603, by way ofthe internal bus 651. The individual co-processors are in the sleepstate before the executions are started, so that little power isdissipated.

FIG. 45 is a diagram showing the internal constructions of the COPA 602.The inside is constructed of two blocks ITFA 700 and EXA 701. The ITFA700 is composed of a command register CMDA 609, a command decoder DEC706, operand registers RA 702, RB 703 and RC 704, and a control circuitCNT 705. The EXA 701 has a proprietary execution unit for processing themultiplications, the divisions and the calculations of roots andabsolute values at a high speed, and a control circuit for controllingthe execution unit. The command sent from the CPU 601 through theinternal bus 651 is latched in the CMDA 609 and decoded by the DEC 706to cause the EXA 701 to execute the operations instructed by thatcommand. The command is divided into one of multiplication, division,and calculations of either roots or absolute values. The source operandsare sent from the CPU 601 and stored in the RA 702 and RB 703, and theexecuted results of the EXA 701 are stored in the RC 704 and read out bythe CPU 601. The EXA 701 is in the sleep state while it is notexecuting. When the command is decoded by the DEC 706, a signal forcausing the EXA 701 to execute the operation instructed by the commandis generated so that the EXA 701 starts its execution. During thisexecution, the EXA 701 is in the active state. After this execution, theEXA 701 stores the results in the RC 704 to clear the CMDA 609 to zero.When the DEC 706 detects that the content of the CMDA 609 is zero, theEXA 701 is brought into the sleep state by asserting a SLEEP signal 707.The CNT 705 controls the reading/writing and zero-clearing operationsfor the individual registers 609, 702, 703 and 704. The ITFA 700 isalways in the active state so as to receive the commands from the CPU atall times. The clock signal 653 generated by the CPG 606 is used in theITFA 700. On the other hand, a EXA clock signal 710 is outputted througha gate circuit 709 and used as a clock for the EXA 701. When the SLEEP707 is asserted, the gate circuit 709 interrupts the EXA clock 710 sothat the clock is not supplied to the EXA 701. As a result, the clock ofthe EXA 701 is also interrupted in the sleep state. By this SLEEPsignal, for example, the switches of the circuits of FIG. 18 to 26 orFIGS. 28 to 32 are controlled to reduce the subthreshold current in thesleep state.

The EXA 701 contains not only a proprietary execution unit but also aregister for latching the intermediate result of the execution, aregister for latching the execution state, and a latch for controllingthe execution. These registers and latches to be used are exemplified bythe circuits of FIG. 35 and FIG. 37. For the case of the circuit of FIG.35, the internal state of the latch is broken once the sleep state isentered. For the case of the circuit of FIG. 37, on the other hand, theinternal state of the latch is not broken even if the sleep state isentered. As a result, when the active state is restored after the sleepstate is once entered, the temporary executions that were suspended canbe reopened.

The COPB is a co-processor for executing the functional calculations oftrigonometric functions and distances and is given an internal structureand an operation similar to that of the COPA 602.

FIG. 46 shows the internal structure of the LM 604. MEM 901 is a memoryunit for storing information such as instructions/data. MCNT 902receives an access demand from the CPU 601 to read the data stored inthe MEM 901 and to write the data in the MEM 901. When an access demandis made by the CPU 601, the MCNT 902 asserts a signal ACT 903 foractivating the MEM 901 and activates the MEM 901. Without the accessdemand, the ACT 903 is negated so that the MEM 901 is in the sleepstate. By this ACT signal, the switches of the circuits of FIGS. 18 to26 or FIGS. 28 to 32, for example, are controlled to reduce thesubthreshold current in the sleep state. In this state, too, the dataare retained in the memory. The MCNT 902 is always in the active stateso that it may receive the access demand from the CPU at all times.

The CPU 601 accesses the instructions or data to the MS 606 through theinternal bus 651, the BUSC 605 and the external bus 652. Only at thistime, the BUSC 605 is activated. FIG. 47 shows the internal structure ofthe BUSC 605. BCNT 800 is a circuit for controlling the access to theexternal bus 652 in response to the demand of the CPU 601. OUTB 801 is adriver circuit for driving the external bus 652, when data are suppliedfrom the internal bus 651 to the external bus 652, and is activated onlyat this time. INB 802 is a driver circuit for driving the internal bus651, when data are supplied from the external bus 652 to the internalbus 651, and is activated only at this time. When the BCNT 800 receivesa write demand for the MS 606 or the IO 607 outside of the chip from theCPU 601, it asserts ACTW 803 to activate the OUTB 801. On the otherhand, when the BCNT receives a read demand of the MS 606 or the IO 607outside of the chip from the CPU 601, it asserts ACTR 804 to activatethe INB 802. For the period other than those operations, the OUTB 801and the INB 802 are in the sleep state. The BCNT 800 is always active sothat it may receive the access demand to the outside of the chip. TheBCNT 800 further outputs the active support signal 654 for the MS 606and the active instruct signal 655 for the IO 607. In case the CPU 601demands the BCNT 800 to access the MS 606, the BCNT 800 detects thedemand to assert the signal 654 to activate the MS 606. The signal 655is also used for similar operations.

The OUTB 801 is exemplified by the output buffer of FIG. 25, and theswitches S_(S) and S_(C) are controlled in accordance with the ACTWsignal. This OUTB drives a high load (e.g., the external bus 652) andneeds the MOS transistors having a large channel width in the number ofthe bus width (e.g., 64 bits) so that its total channel width isremarkably large. As a result, the reduction of the subthreshold currentof the OUTB highly contributes to the reduction of the current of theentire system.

The INB 802 used is exemplified by the input buffer of FIG. 26, and theACTR signal is supplied to the SB terminal. As a result, the internalbus 651 has its voltage level fixed in the sleep state. Thus, the unitsCOPA, COPB and LM connected with the internal bus 651 can be exemplifiedby the circuits of FIGS. 18 to 25 so that these units can have theirsubthreshold currents reduced easily.

The MS 606 is exemplified by a DRAM. This DRAM may be an ordinary DRAMor a synchronous DRAM, as is disclosed on pp. 43-49 of IEEE Spectrum,October 1992. In this synchronous DRAM, the supply of the clock to thechip inside can be controlled by a clock enable/disable signal so thatthe current dissipation can be effectively reduced by utilizing thesignal. Specifically, the clock supply to the chip inside is interruptedin the sleep state. Moreover, the subthreshold current of the internalcircuit can be reduced by using the circuit of FIG. 26 as the inputbuffer of the synchronous DRAM and by applying the clock enable/disablesignal to the SB terminal.

FIG. 48 illustrates examples of the overall operations of themicroprocessor 600. The abscissa indicates the time, and hatching linesindicate that the individual units or blocks are active. In thisexample, the CPU 601 issues a dividing command to the COPA 602 at timeT1, and the COPA 602 executes the divisions from time T1 to time T2 andreports the end of calculations to the CPU 601 at time T2 until it comesagain into the sleep state. After this, the CPU 601 issues a distancecalculating command to the COPB 603 at time T3, and the COPB 603executes the distance calculations from time T3 to time T4 and reportsthe end of calculations to the CPU 601 at time T4 until it comes againinto the sleep state. The LM 604 is activated only when the access todata is demanded by the CPU 601. The BUSC 605 is also activated onlywhen the CPU 601 accesses the outside. Thus, the individual units andblocks in the microprocessor 600 have their active/sleep statesprecisely controlled so that the microprocessor 600 can have its powerdissipation greatly reduced.

This embodiment of the invention is applied internally of the chip, butit is self-explanatory that this invention can be expanded to anembodiment of a computer system constructed of a plurality of chips. Forexample, this invention can be easily applied to the case in which theindividual units 601 to 605 of FIG. 44 are made of different chips.

Respective embodiments of the present invention have been describedabove taking a MOS semiconductor circuit device as an example, but thepresent invention is also applicable to a metal insulator semiconductor(MIS) circuit device.

We claim:
 1. A semiconductor integrated circuit device, comprising:afirst logic gate having a first MOSFET whose source/drain path isbetween a first node and a first output node; a second logic gate havinga second MOSFET whose source/drain path is between a second node and asecond output node; a first control circuit connected to receive acontrol signal, said first control circuit being provided between saidfirst node and a first potential point, wherein said first controlcircuit permits a first current to flow between said first node and athird node in response to said control signal being in a first state,said first control circuit limits the current flowing between said firstnode and said third node to a current smaller than said first current inresponse to said control signal being set to a second state differentfrom said first state, and a second control circuit provided betweensaid second node and a second potential point, wherein said secondcontrol circuit permits a second current to flow between said secondnode and a fourth node in response to said control signal being in saidfirst state, said second control circuit limits the current flowing insaid second logic gate to a current smaller than said second current inresponse to said control signal being in said second state, a thirdlogic gate having a fifth MOSFET whose source/drain path is between afifth node and a third output node and a sixth MOSFET whose source/drainpath is between a sixth node and said third output node, and a fourthlogic gate having a seventh MOSFET whose source/drain path is between aseventh node and a fourth output node and an eighth MOSFET whosesource/drain path is between an eighth node and said fourth output node;wherein said first logic gate has a third MOSFET whose source/drain pathis between said third node and said first output node; wherein saidsecond logic gate has a fourth MOSFET whose source/drain path is betweensaid fourth node and said second output node; wherein said third node isdirectly coupled to said second potential point; wherein said fourthnode is directly coupled to said first potential point; wherein saidfifth node is connected to said first control circuit; wherein saidseventh node is connected to said second control circuit; wherein saidsixth node is directly coupled to said second potential point; andwherein said eighth node is directly coupled to said first potentialpoint.
 2. A semiconductor integrated circuit device according to claim1,wherein the gate of said second MOSFET is coupled to said first outputnode; wherein the gate of said fifth MOSFET is coupled to said secondoutput node; wherein the gate of said seventh MOSFET is coupled to saidthird output node; and wherein said first, second, fifth and seventhMOSFETs are in an off state when said control signal is in said secondstate.
 3. A semiconductor integrated circuit device according to claim2,wherein said first control circuit has a first switch between saidfirst node and said first potential point; and wherein said secondcontrol circuit has a second switch between said second node and saidsecond potential point.
 4. A semiconductor integrated circuit deviceaccording to claim 3,wherein said first control circuit has a firstresistor between said first node and said first potential point inparallel connection to said first switch; and wherein said secondcontrol circuit has a second resistor in parallel connection to saidsecond switch between said second node and said second potential point.5. A semiconductor integrated circuit device according to claim3,wherein each of said first and second switches is a MOSFET whoseabsolute value of threshold voltage is larger than an absolute value ofthreshold voltage of each of said first, second, third, forth, fifth,sixth, seventh and eighth MOSFETs, respectively.
 6. A semiconductorintegrated circuit, comprising:a first group of first logic gates eachhaving at least a first MOSFET; a first control circuit connected toreceive a control signal; a second group of second logic gates eachhaving at least a second MOSFET; wherein each source/drain path of saidfirst MOSFETs of said first logic gates is coupled between a first nodeand a second node; wherein said first control circuit is providedbetween said first node and a first potential point, wherein said firstcontrol circuit permits a first current to flow in said source/drainpaths of the said first MOSFETs in response to said control signal beingin a first state; wherein said first control circuit limits the currentflowing in said source/drain paths of said first MOSFETs to a currentsmaller than said first current in response to said control signal beingset to a second state different from said first state; wherein outputnodes of said first logic gates have a same first logic level when saidcontrol signal is in said second state; wherein output nodes of saidsecond logic gates have a same second logic level different than thefirst logic level of said first logic gates when said control signal isin said second state; and wherein said second node is directly coupledto a second potential point.
 7. A semiconductor integrated circuitaccording to claim 6,wherein said first logic level is lower than saidsecond logic level; and wherein said first potential point is higherthan said second potential point.
 8. A semiconductor integrated circuitaccording to claim 6,wherein said first logic level is higher than saidsecond logic level; and wherein said first potential point is lower thansaid second potential point.
 9. A semiconductor integrated circuitaccording to claim 6, further comprising:a second control circuitprovided between a fourth node and said second potential point, whereineach source/drain path of said second MOSFETs of said second logic gatesis coupled between a third node and said fourth node; wherein saidsecond control circuit permits a second current to flow in saidsource/drain paths of said second MOSFETs in response to said controlsignal being in a first state; wherein said second control circuitlimits the current flowing in said source/drain paths of said secondMOSFETs to a current smaller than said second current in response tosaid control signal being in a second state different from said firststate; wherein said third node is directly coupled to said firstpotential point; wherein said first logic level is lower than saidsecond logic level; and wherein said first potential point is higherthan said second potential point.
 10. A semiconductor integrated circuitdevice according to claim 6,wherein said first control circuit has athird MOSFET between said first node and said first potential point. 11.A semiconductor integrated circuit device according to claim 10,whereinsaid first control circuit has a first resistor between said first nodeand said first potential point in parallel to said third MOSFET.
 12. Asemiconductor integrated circuit device according to claim 6,whereinsaid control circuit not only limits but stops the current flowing insaid source/drain paths of the said first MOSFET.
 13. A semiconductorintegrated circuit, comprising:a first group of first logic gates eachhaving at least a first MOSFET, a first control circuit connected toreceive a control signal; a second group of second logic gates eachhaving at least a second MOSFET; wherein each source/drain path of saidfirst MOSFETs of said first logic gates is coupled between a first nodeand a second node; wherein said first control circuit is providedbetween said first node and a first potential point, wherein said firstcontrol circuit permits a first current to flow in said first logicgates in response to said control signal being in a first state; whereinsaid first control circuit limits the current flowing in said firstlogic gates to a current smaller than said first current in response tosaid control signal being set to a second state different from saidfirst state; wherein output nodes of said first logic gates have a samefirst logic level when said control signal is in said second state;wherein output nodes of said second logic gates have a same second logiclevel different than the first logic level of said first logic gateswhen said control signal is in said second state; and wherein saidsecond node is directly coupled to a second potential point.